Dynamic signal detection threshold

ABSTRACT

A method for setting a signal detection threshold according to one embodiment includes determining a measure of a noise floor in a signal derived from a radio frequency signal received by an antenna using a same circuit used to detect a subcarrier signal during transmitting and prior to sending a command to a transponder to respond; and setting a signal detection threshold above the noise floor. Such methodology may also be implemented as a system using logic for performing the various operations. Additional systems and methods are also presented.

RELATED APPLICATIONS

This application claims priority to U.S. Provisional Patent ApplicationNo. 61/360,883, filed Jul. 1, 2010, and which is herein incorporated byreference.

BACKGROUND

The use of Radio Frequency Identification (RFID) tags are quicklygaining popularity for use in the monitoring and tracking of an item.RFID technology allows a user to remotely store and retrieve data inconnection with an item utilizing a small, unobtrusive tag. As an RFIDtag operates in the radio frequency (RF) portion of the electromagneticspectrum, an electromagnetic or electrostatic coupling can occur betweenan RFID tag affixed to an item and an RFID tag reader. This coupling isadvantageous, as it precludes the need for a direct contact or line ofsight connection between the tag and the reader.

In some currently used passive and semi-passive RFID tags, during the‘read’ cycle, the reader generally transmits a continuous unmodulatedcarrier signal. A distant RFID tag includes a RF switch connected to thetag's antenna, which repetitively alternates its state at a rate calledthe ‘backscatter link frequency’ (BLF). This RF switch effectivelymodulates the carrier signal in the tag received from the transmitter,creating sidebands surrounding the carrier frequency, and separated fromthe carrier frequency by the backscatter link frequency. For example, ifthe carrier frequency is 900 MHz and the tag backscatter modulation isat 160 KHz, side bands present in the return signal are about 900MHz+˜160 KHz and 900 MHz+˜160 KHz. These sidebands are re-radiated bythe tag's antenna, and are recovered by the reader, e.g., the readerdetects and demodulates one or both of the side bands to obtain the datareturned by the tag.

The above description is one typical way in which the tag communicatesinformation to the reader. The tag does not create RF power, but insteadmodulates incoming RF power from the reader's transmitter, and in sodoing, converts some of that incoming power to sideband frequencieswhich can be separately recovered by the reader. These backscattersidebands only exist when (and because) the reader is transmitting.

The current RFID standard includes a wide range of reverse linkparameters such as backscatter link frequency and data rates to provideflexibility in various applications. Most readers are designed to usespecific parameter combinations at which the demodulator operates, i.e.,the readers are designed to operate at a specific frequency and datarate. Such designs have traditionally been favored due to theirsimplicity, lower cost, and effectiveness at talking to particular tagdesigns. Moreover, in long distance communications with semi-passive andactive tags, the low return signal strength urges towards settingsdirected to a specific frequency. However, such implementations areunable to cover a continuum of reverse link backscatter link frequenciesand data rates.

SUMMARY OF THE INVENTION

A method for setting a signal detection threshold according to oneembodiment includes determining a measure of a noise floor in a signalderived from a radio frequency signal received by an antenna using asame circuit used to detect a subcarrier signal during transmitting andprior to sending a command to a transponder to respond; and setting asignal detection threshold above the noise floor.

Such methodology may also be implemented as a system using logic forperforming the various operations.

Any of these embodiments may be implemented in an RFID system, which mayinclude an RFID tag and/or reader.

Other aspects, advantages and embodiments of the present invention willbecome apparent from the following detailed description, which, whentaken in conjunction with the drawings, illustrate by way of example theprinciples of the invention.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a system diagram of an RFID system according to oneembodiment.

FIG. 2 is a diagram depicting a circuit layout of an integrated circuit(IC) and various control circuitry according to an illustrativeembodiment for implementation in an RFID tag.

FIG. 3 is a diagram of an RFID reader circuit according to oneembodiment.

FIG. 4 is a system diagram of an illustrative demodulator according toone embodiment.

FIG. 5 is a system diagram of the derotator of FIG. 4.

FIG. 6 is a system diagram of the frequency error filter module of FIG.5.

FIG. 7 is a system diagram of the NCO of FIG. 5.

FIG. 8 is a system diagram of the subcarrier detector module of FIG. 4.

FIG. 9 is a system diagram of the frequency detector module of FIG. 4.

FIG. 10 is a system diagram of the magnitude normalization module ofFIG. 9.

FIG. 11 is a system diagram of the quadrature correlator FM demodulatorof FIG. 9

FIG. 12 is a system diagram of the preintegrator module of FIG. 4.

FIG. 13 is a system diagram of one preintegrator of FIG. 12.

FIG. 14 is a system diagram of the correlator module of FIG. 4.

FIG. 15 is a series of charts depicting illustrative outputs of thecorrelator.

FIG. 16 is a system diagram of one windowed integrator of FIG. 14.

FIG. 17 is a system diagram of one variable delay module of FIG. 14.

FIG. 18 is a system diagram of the preamble delimiter detection moduleof FIG. 4.

FIG. 19 is a system diagram of the state machine of FIG. 18.

FIG. 20 is a system diagram of the state sequence recognition module ofFIG. 18.

FIG. 21 is a system diagram of the clock recovery module of FIG. 4.

FIG. 22 is a system diagram of the loop filter of FIG. 21.

FIG. 23 is a system diagram of the end of modulation detection module ofFIG. 4.

FIG. 24 is a system diagram of a 3-point window integrator of FIG. 23.

FIG. 25 is a system diagram of the leaky integrator of FIG. 23.

FIG. 26 is a system diagram of the automatic gain control (AGC) moduleof FIG. 4.

FIG. 27 is a system diagram of the received signal strength indicator(RSSI) module of FIG. 4.

FIG. 28 depicts a process flow for a method for demodulating a radiofrequency signal, according to one general embodiment.

FIG. 29 is a series of charts showing signals according to oneembodiment.

FIG. 30 depicts a process flow for a method for jam setting an initialfrequency of a data clock recovery loop, according to one generalembodiment.

FIG. 31 depicts a process flow for a method for processing a signalderived from a radio frequency signal at some rate in a range ofallowable data rates, according to one general embodiment.

FIG. 32 depicts a process flow for a method for demodulating a datasignal, according to one general embodiment.

FIG. 33 is a series of charts showing signals according to oneembodiment.

FIG. 34 depicts a process flow for a method for processing basebandsignals by reducing the number of samples required to be processed,according to one general embodiment.

FIG. 35 depicts a process flow for a method for subcarrierdownconversion and data detection, according to one general embodiment.

FIG. 36 depicts a process flow for a method for detecting a pattern in asignal, according to one general embodiment.

FIG. 37 depicts a signal after slicing and filtering according to oneembodiment.

FIG. 38 is a table of illustrative values according to one embodiment.

FIG. 39 is a graphical depiction of an illustrative relational tableusable in a fuzzy logic approach according to one embodiment.

FIG. 40 depicts a process flow for a method for detecting an end ofmodulation in a backscattered radio frequency signal, according to onegeneral embodiment.

FIG. 41 depicts a process flow for a method for estimating a strength ofa radio frequency signal, according to one general embodiment.

FIG. 42 depicts a process flow for a method for setting a signaldetection threshold, according to one general embodiment.

DETAILED DESCRIPTION

The following description is made for the purpose of illustrating thegeneral principles of the present invention and is not meant to limitthe inventive concepts claimed herein. Further, particular featuresdescribed herein can be used in combination with other describedfeatures in each of the various possible combinations and permutations.

Unless otherwise specifically defined herein, all terms are to be giventheir broadest possible interpretation including meanings implied fromthe specification as well as meanings understood by those skilled in theart and/or as defined in dictionaries, treatises, etc.

It must also be noted that, as used in the specification and theappended claims, the singular forms (e.g., “a,” “an” and “the”) includeplural referents unless otherwise specified.

In the drawings, like and equivalent elements are numbered the samethroughout the various figures.

FIG. 1 depicts an RFID system 100 according to one of the variousembodiments, which may include some or all of the following componentsand/or other components. As shown in FIG. 1, one or more RFID tags 102are present. Each RFID tag 102 in this embodiment includes a controllerand memory, which are preferably embodied on a single chip as describedbelow, but may also or alternatively include a different type ofcontroller, such as an application specific integrated circuit (ASIC),processor, an external memory module, etc. For purposes of the presentdiscussion, the RFID tags 102 will be described as including a chip.Each RFID tag 102 may further include or be coupled to an antenna 105.

An illustrative chip is disclosed below, though actual implementationsmay vary depending on how the tag is to be used. In general terms, apreferred chip includes one or more of a power supply circuit to extractand regulate power from the RF reader signal; a detector to decodesignals from the reader; a backscatter modulator, an interface to atransmitter to send data back to the reader; anti-collision protocolcircuits; and at least enough memory to store its unique identificationcode, e.g., Electronic Product Code (EPC).

While RFID tags 102, according to some embodiments, are functional RFIDtags, other types of RFID tags 102 include merely a controller withon-board memory, a controller and external memory, etc.

Each of the RFID tags 102 may be coupled to an object or item, such asan article of manufacture, a container, a device, a person, etc.

With continued reference to FIG. 1, a remote device 104, such as aninterrogator or “reader,” communicates with the RFID tags 102 via an airinterface, preferably using standard RFID protocols. An “air interface”refers to any type of wireless communications mechanism, such as theradio-frequency signal between the RFID tag and the remote device(reader). The RFID tag 102 executes the computer commands that the RFIDtag 102 receives from the reader 104.

The system 100 may also include an optional backend system such as aserver 106, which may include databases containing information and/orinstructions relating to RFID tags and/or tagged items.

As noted above, each RFID tag 102 may be associated with a uniqueidentifier. Such identifier is preferably an EPC code. The EPC is asimple, compact identifier that uniquely identifies objects (items,cases, pallets, locations, etc.) in the supply chain. The EPC is builtaround a basic hierarchical idea that can be used to express a widevariety of different, existing numbering systems, like the EAN.UCCSystem Keys, UID, VIN, and other numbering systems. Like many currentnumbering schemes used in commerce, the EPC is divided into numbers thatidentify the manufacturer and product type. In addition, the EPC uses anextra set of digits, a serial number, to identify unique items. Atypical EPC number contains:

-   -   1. Header, which identifies the length, type, structure, version        and generation of EPC;    -   2. Manager Number, which identifies the company or company        entity;    -   3. Object Class, similar to a stock keeping unit or SKU; and    -   4. Serial Number, which is the specific instance of the Object        Class being tagged. Additional fields may also be used as part        of the EPC in order to properly encode and decode information        from different numbering systems into their native        (human-readable) forms.

Each RFID tag 102 may also store information about the item to whichcoupled, including but not limited to a name or type of item, serialnumber of the item, date of manufacture, place of manufacture, owneridentification, origin and/or destination information, expiration date,composition, information relating to or assigned by governmentalagencies and regulations, etc. Furthermore, data relating to an item canbe stored in one or more databases linked to the RFID tag. Thesedatabases do not reside on the tag, but rather are linked to the tagthrough a unique identifier(s) or reference key(s).

RFID systems may use reflected or “backscattered” radio frequency (RF)waves to transmit information from the RFID tag 102 to the remote device104, e.g., reader. Since passive (Class-1 and Class-2) tags get all oftheir power from the reader signal, the tags are only powered when inthe beam of the reader 104.

The Auto ID Center EPC-Compliant tag classes are set forth below:

Class-1

-   -   Identity tags (RF user programmable, range ˜3 m)    -   Lowest cost

Class-2

-   -   Memory tags (20 bit address space programmable at ˜3 m range)    -   Security & privacy protection    -   Low cost

Class-3

-   -   Semi-passive tags (also called semi-active tags and battery        assisted passive (BAP) tags)    -   Battery tags (256 bits to 2M words)    -   Self-Powered Backscatter (internal clock, sensor interface        support)    -   ˜100 meter range    -   Moderate cost

Class-4

-   -   Active tags    -   Active transmission (permits tag-speaks-first operating modes)    -   ˜300 to ˜1,000 meter range    -   Higher cost

In RFID systems where passive receivers (i.e., Class-1 and Class-2 tags)are able to capture enough energy from the transmitted RF to power thetag, no batteries are necessary. In systems where distance preventspowering a tag in this manner, an alternative power source must be used.For these “alternate” systems (e.g., semi-active, semi-passive orbattery-assisted), batteries are the most common form of power. Thisgreatly increases read range, and the reliability of tag reads, becausethe tag does not need power from the reader to respond. Class-3 tagsonly need a 5 mV signal from the reader in comparison to the 500 mV thatClass-1 and Class-2 tags typically need to operate. This 100:1 reductionin power requirement along with the reader's ability to sense a verysmall backscattered signal permits Class-3 tags to operate out to a freespace distance of 100 meters or more compared with a Class-1 range ofonly about 3 meters. Note that semi-passive and active tags with builtin passive mode may also operate in passive mode, using only energycaptured from an incoming RF signal to operate and respond, at a shorterdistance up to 3 meters.

Active, semi-passive and passive RFID tags may operate within variousregions of the radio frequency spectrum. Low-frequency (30 KHz to 500KHz) tags have low system costs and are limited to short reading ranges.Low frequency tags may be used in security access and animalidentification applications for example. Ultra high-frequency (860 MHzto 960 MHz and 2.4 GHz to 2.5 GHz) tags offer increased read ranges andhigh reading speeds.

A basic RFID communication between an RFID tag and a remote device,e.g., reader, typically begins with the remote device, e.g., reader,sending out signals via radio wave to find a particular RFID tag viasingulation or any other method known in the art. The radio wave hitsthe RFID tag, and the RFID tag recognizes the remote device's signal andmay respond thereto. Such response may include exiting a hibernationstate, sending a reply, storing data, etc.

Embodiments of the RFID tag are preferably implemented in conjunctionwith a Class-3 or higher Class IC chip, which typically contains theprocessing and control circuitry for most if not all tag operations.FIG. 2 depicts a circuit layout of a Class-3 IC 200 and the variouscontrol circuitry according to an illustrative embodiment forimplementation in an RFID tag 102. It should be kept in mind that thepresent invention can be implemented using any type of RFID tag, and thecircuit 200 is presented as only one possible implementation.

The Class-3 IC of FIG. 2 can form the core of RFID chips appropriate formany applications such as identification of pallets, cartons,containers, vehicles, or anything where a range of more than 2-3 metersis desired. As shown, the chip 200 includes several circuits including apower generation and regulation circuit 202, a digital command decoderand control circuit 204, a sensor interface module 206, a C1G2 interfaceprotocol circuit 208, and a power source (battery) 210. A display drivermodule 212 can be added to drive a display.

A forward link AM decoder 216 uses a simplified phase-lock-looposcillator that requires only a small amount of chip area. Preferably,the circuit 216 requires only a minimum string of reference pulses.

A backscatter modulator block 218 preferably increases the backscattermodulation depth to more than 50%.

A memory cell, e.g., EEPROM, is also present, and preferably has acapacity from several kilobytes to one megabyte or more. In oneembodiment, a pure, Fowler-Nordheim direct-tunneling-through-oxidemechanism 220 is present to reduce both the WRITE and ERASE currents toabout 2 μA/cell in the EEPROM memory array. Unlike any RFID tags builtto date, this permits reliable tag operation at maximum range even whenWRITE and ERASE operations are being performed. In other embodiments,the WRITE and ERASE currents may be higher or lower, depending on thetype of memory used and its requirements.

Preferably, the amount of memory available on the chip or otherwise isadequate to store data such that the external device need not be inactive communication with the remote device, e.g., reader.

The module 200 may also incorporate a security encryption circuit 222for operating under one or more security schemes, secret handshakes withreaders, etc.

The RFID tag may have a dedicated power supply, e.g. battery; may drawpower from a power source of the electronic device (e.g., battery, ACadapter, etc.); or both. Further, the RFID tag may include asupplemental power source. Note that while the present descriptionrefers to a “supplemental” power source, the supplemental power sourcemay indeed be the sole device that captures energy from outside the tag,be it from solar, RF, kinetic, etc. energy.

FIG. 3 illustrates an RFID reader circuit 300 according to oneembodiment. As shown, the circuit includes an RF source 320 thatgenerate a carrier signal. Data may be added to the carrier wave via RFmodulation, in a manner that is well known.

In some readers, such as the one shown in FIG. 3, the antennaconfiguration is ‘monostatic’ which means that the sidebands created bythe distant tag are recovered via the same antenna that the reader'stransmitter uses to transmit the carrier signal. In a monostatic reader,a power amplifier 302 is connected to a signal splitter 306, which isconnected to an antenna 310 and a low noise amplifier 304 in the receivepath. In some other readers, the antenna configuration is ‘bistatic’which means that the sidebands created by the distant tag are recoveredby the reader via a separate antenna. A bistatic reader differs from amonostatic reader in that each reader has a power amplifier, low noiseamplifier, and antenna, but in the bistatic reader, the power amplifieris connected directly to a transmit antenna, a separate receive antennaconnects directly to the low noise amplifier, and there may be no signalsplitter.

In the receive path, the incoming signal is received by the antenna 310and passes through the signal splitter 306 to the low noise amplifier.The signal is filtered by a bandpass filter 308, and a sample of thecarrier signal is combined with the incoming signal at a mixer 322 toprovide carrier cancellation. Assuming the carrier cancellation iseffective, the sideband signals remain and are filtered by a secondfilter 324, amplified by an amplifier 326, and converted to a digitalsignal by an analog to digital converter 328. The digital signal is thenprocessed in a digital signal recovery section 330, which may include adigital signal processor (DSP) or other circuitry to process the digitalsignal.

Illustrative carrier signal frequencies correspond to those noted abovein the description of the illustrative RFID tags. By way of example,assume a carrier signal produced by an RFID reader is about 900 MHz.Typically, the RFID tag signal is sent at an offset due to modulation,e.g., 320 KHz, rendering sidebands of 900 MHz±320 KHz. Thus, the RFIDtag signal coming back is not the same as the signal being emitted bythe RFID reader, and can be detected by the reader. However, thereceiver is literally swamped with unwanted signals as well as noisegenerated by the reader itself, making it difficult to discern themodulations of the incoming RFID tag signal from the multitude ofincoming signals and nose.

The following section describes various embodiments that may be embodiedin the digital signal recovery section 330 of FIG. 3, or other type ofreader design. It should be noted, however, that various embodiments ofthe inventive devices and methodologies described herein may be used inconjunction with any type of RF system, and is particularly adapted foruse in long range RFID applications, such as in or as a readercommunicating with semi-passive and active RFID tags. Illustrativereaders in which the device may be implemented or embodied includefixed-location readers, portable readers, handheld “gun-type” readers,etc.

In particularly preferred embodiments, the device implements any ofseveral new technologies to enable communication over a continuum ofreverse link backscatter link frequencies and/or data rates. Thisenables a reader employing or embodying such device to be able toquickly and easily change modes for use in different applications, usewith different types of tags, use in different jurisdictions, etc.

New technology implemented in various embodiments includes:

-   -   Subcarrier frequency acquisition    -   Data clock recovery loop jam set using subcarrier frequency        estimate    -   Variable length correlator    -   Half bit correlator combining for Miller data demodulation    -   Baseband preintegrator to reduce computation requirements at        lower data rates    -   Dual complex subcarrier downconversion    -   Variable rate preamble detection using ratiometric methods.    -   End of modulation detection    -   RSSI estimate on variable length correlator output    -   Dynamic signal detection threshold

Each of these technologies is described in turn. It should be understoodthat some embodiments may implement only one technology from the list,while other embodiments may implement more than one. Moreover, asdiscussed in more detail below, a technology may be implemented as apure method; in hardware or software logic such as hardware (circuit,ASIC, processor, etc.), reconfigurable logic (e.g., FPGA), software,firmware, etc.; modules of such logic; and combinations thereof. In someapproaches, the logic transforms a signal to another state, and/oreffects a transformation of data into another usable form, any of whichmay be usable in a further operation and/or by other logic.

As will soon become apparent, some embodiments of devices implementingmethodology presented herein may communicate with a variety of RFID tagsand other RF devices over a broad spectrum of frequencies and data ratesin a single device. To illustrate, and presented by way of example only,a reader according to one embodiment is able to communicate atsubstantially all bit rates in a range of about 2.5 Kbits/second toabout 160 Kbits/second, and at all backscatter link frequencies in arange of about 25 KHz to about 640 KHz, where “about X” Kbits/second,“about X” KHz, etc. means “X±10%.”

To place the methodologies in a context, a general system design of anillustrative demodulator is first presented to show one of the manypossible implementations and permutations in which embodiments of theinvention may be embodied. It is again stressed that the illustrativesystem design is presented by way of example only to show one of themany possible implementations and permutations that will become apparentto one skilled in the art upon reading this specification.

FIG. 4 is a system diagram of an illustrative demodulator 400 accordingto one embodiment. The demodulator 400 may form part of, or be, a gatearray instantiation of a digital signal processing block, according topreferred embodiments. The illustrative demodulator 400 includes aderotator 402 having quadrature I and Q inputs 404, 406, respectively,from the complex RF downconverter of the host device receiver, which maybe an RFID reader or other device. In the RFID reader implementation,the I and Q inputs are received at baseband from the receive path of thehost device.

FIG. 5 is a system diagram of the derotator 402 of FIG. 4. Note that anytype of derotator known in the art may be used, and the circuit shown ispresented by way of example only to show one possible implementation.

With continued reference to FIG. 5, an eight phase derotator is shown.Inputs include the I and Q inputs 404, 406; a frequency error signalinput 502; a subcarrier detect signal input 504; and a reset signalinput 506 which receives a reset signal. The signals and their originsare described in more detail below. A frequency error filter module 507is coupled to the frequency error signal input 502. The frequency errorfilter module 507 is initiated by a signal from a subcarrier detectenable module 509, that is in turn coupled to the subcarrier detectsignal input 504. The frequency error filter module 507, when enabled,generates a numerically controlled oscillator (NCO) input signal basedon the frequency error signal. The NCO input signal is sent to an NCO511, the output of which is used to generate cosine and sine signals foruse by a positive subcarrier derotator 508 and a negative subcarrierderotator 510.

Outputs of the subcarrier derotators 508, 510 include baseband I and Qoutputs 512, 514; negative baseband I and Q outputs 516, 518; and afrequency offset output 520.

FIG. 6 is a system diagram of the frequency error filter module 507 ofFIG. 5. Note that any type of frequency error filter module or suitablealternative known in the art may be used, and the circuit shown ispresented by way of example only to show one possible implementation.With continued reference to FIG. 6, the frequency error is received andscaled to generate the NCO input signal.

FIG. 7 is a system diagram of the NCO 511 of FIG. 5. Note that any typeof NCO or suitable alternative known in the art may be used, and thecircuit shown is presented by way of example only to show one possibleimplementation. With continued reference to FIG. 7, the NCO inputsignal, which is the processed frequency error signal, is received bythe NCO, which in the implementation shown, includes a modulo N-counter(counts up to N), where N is set by the NCO_mod N parameter. As shown, aloop increment is applied, so in the absence of any error, the NCOoperates at a predefined frequency that is defined by the two parametersNCO loop increment (NCO_loop_incr) and NCO_mod N.

As noted in FIG. 4, the subcarrier detect signal is generated by asubcarrier detector module 408. FIG. 8 is a system diagram of thesubcarrier detector module 408 of FIG. 4. Note that any type ofsubcarrier detector module or suitable alternative known in the art maybe used, and the circuit shown is presented by way of example only toshow one possible implementation. The subcarrier detector module 408performs a scaling on filtered I and Q baseband signals (ss10I, ss10Q)output by the derotator (402, FIG. 4) using a scaler module 802. Thescaler module 802 is programmable from configuration parameters appliedduring build, on the fly, etc. A complex to magnitude module 804receives the output of the scaler module 802 and generates a magnitude(Mag) signal, which is input to a leaky integrator 806. The output ofthe leaky integrator 806 is a subcarrier power detect signal. This inturn is used to detect when the subcarrier signal level has exceeded adefinable or predefined threshold set above a definable or predefinednoise floor. When the subcarrier power detect signal (sc_pwr_det)exceeds the noise floor or some predetermined or dynamically determinedthreshold (e.g., a threshold slightly above the noise floor), thesubcarrier detect signal (sc_detect, FIG. 4) indicates presence of thesubcarrier and may be used by the derotator 402 (FIG. 4) as mentionedabove.

As noted in FIG. 4, the frequency error signal is generated by afrequency detector module 410. FIG. 9 is a system diagram of thefrequency detector module 410 of FIG. 4. Note that any type of frequencydetector module or suitable alternative known in the art may be used,and the circuit shown is presented by way of example only to show onepossible implementation. A prescaler module 902 receives the filtered Iand Q baseband signals (ss10I, ss10Q) and outputs scaled versionsthereof to a magnitude normalization module 904 and a quadraturecorrelator 906. The prescaler may be configurable from configurationparameters applied during build, on the fly, etc.

FIG. 10 is a system diagram of the magnitude normalization module 904 ofFIG. 9. Note that any type of magnitude normalization module or suitablealternative known in the art may be used, and the circuit shown ispresented by way of example only to show one possible implementation. Asshown, the illustrative magnitude normalization module 904 includesmultipliers 1002, 1004 for each of the scaled I and Q signals, and anadder 1006.

FIG. 11 is a system diagram of the quadrature correlator FM demodulator906 of FIG. 9, also referred to herein as a quadrature correlator FMdiscriminator. Note that any type of quadrature correlator FMdemodulator or suitable alternative known in the art may be used, andthe circuit shown is presented by way of example only to show onepossible implementation. As shown, the illustrative quadraturecorrelator FM demodulator 906 includes multipliers 1102, 1104, and delaymodules 1108, 1110, where the delay modules shown have a 2 sample delay,but other approaches may have a 1, 3, 5, etc. sample delay. An adder1106 receiving outputs of the multipliers 1102, 1104. In the embodimentshown, the adder adds: QI′-IQ′, where ′ (prime) denotes a delayedversion of the signal. Particularly, the signals are multiplied andsubtracted to produce an output that is proportional to the frequencyerror.

If the incoming signal is large, the frequency error appears largebecause it is proportional to the signal amplitude. But if this isnormalized by the amplitude of the signal, the frequency error does notappear large. Thus, referring again to FIG. 9, the magnitudenormalization module 904 provides the scaling that ultimately is used togenerate a proper frequency error signal. This design acts as aquadrature correlator frequency modulation (FM) discriminator withmagnitude normalization. Moreover, because tags typically do not have agood clock frequency reference, this approach allows reading of thesignal no matter what state the subcarrier is in, and in spite of noise.

Referring again to FIG. 9, a frequency error generator module 908receives the outputs from the magnitude normalization module 904 and thequadrature correlator FM demodulator 906 and generates a frequency errorsignal. In the example shown, the frequency error generator module 908includes a right shifter which normalizes the data. The quadraturecorrelator 906 in the configuration shown in FIG. 11 is a frequencydiscriminator whose output is proportional to the frequency error, andis normalized by magnitude squared in the frequency error generatormodule 908. The result is that the output (freq_error) of the frequencyerror generator module 908 is not a function of magnitude, but rather offrequency error regardless of whether the incoming signal is small orlarge. The output of the frequency detector module 410 or derivativethereof may be used by the derotator 402 (FIG. 4) as described above.

Referring again to FIG. 4, the baseband I and Q outputs from thederotator are received by a preintegrator module 412. FIG. 12 is asystem diagram of the preintegrator module 412 of FIG. 4. Note that anytype of preintegrator module or suitable alternative known in the artmay be used, and the circuit shown is presented by way of example onlyto show one possible implementation. As shown, the illustrativepreintegrator module 412 includes a top section having an automatic gaincontrol (AGC) section 1202, which may be a right shifter acting as anautomatic gain control, and two preintegrators 1204, 1206, one for I andone for Q. In some embodiments, only this section may be present. Inother embodiments, and as shown, two processing paths may be providedfor the positive and negative baseband signals, e.g., the top section isused for the positive subcarrier signal, and the AGC and preintegratorare duplicated for the negative subcarrier signal in the lower section.

The preintegrator module 412 selectively reduces the number of samplesin the baseband signals being processed by the device. At lower datarates, when there are more than enough samples per symbol, the samplesmay be preintegrated so that the system does not have to processsignificantly more samples than are needed for accurate data recovery.When to reduce the number of samples may be dependent upon a predefinedthreshold corresponding, e.g., to a data rate; may be derived from atable; input from another module or controller; etc.

FIG. 13 is a system diagram of one preintegrator 1204 of FIG. 12. Notethat any type of preintegrator or suitable alternative known in the artmay be used, and the circuit shown is presented by way of example onlyto show one possible implementation.

Referring again to FIG. 4, the outputs from the preintegrator module 412are received by a correlator module 414. FIG. 14 is a system diagram ofthe correlator module 414 of FIG. 4. Note that any type of correlatormodule or suitable alternative known in the art may be used, and thecircuit shown is presented by way of example only to show one possibleimplementation. As shown, the illustrative correlator module 414 has onesection (upper sideband section) for the positive subcarrier signals,and another section (lower sideband section) for the negative subcarriersignals. In the embodiment shown, both sections may be used tocoherently combine both subcarrier sideband signals. In anotherembodiment, the upper or lower sideband sections may be used at a giventime to avoid interference. In yet another embodiment, the system mayselect to use the output of the upper sideband section, the lowersideband section, or both sideband sections for a given operation. Whichsection or section to choose may be determined based on interference inthe signal, whether a stronger signal is desired, e.g. to improvedownconversion, to avoid zero beat distortion caused by Dopplerfrequency shifts from moving tags, etc.

Referring to FIG. 14, the correlator module 414 employs a series ofcorrelator blocks 1402, which in some embodiments may be windowedintegrators that are each a half bit, half symbol integrator correlator,and are each configurable for different lengths thereby allowing thesystem to handle a large variety of data lengths. A correlator lengthselect signal may be used to set the length configuration of thecorrelator blocks 1402. Pairs of correlator variable delay modules 1404provide a half of a symbol delay into two of the correlator blocks 1402(I and Q) in each section. The result is that each section has twonondelayed integrators and two delayed integrators.

The subcarrier combine section 1406 in the embodiment shown isparticularly advantageous in that the signals are kept separate untilthe output of the complex to magnitude modules 1408 so that any Dopplershift due to fast moving tags does not cause any beat note problems.Moreover, combining the sideband signal after the correlators allowsimproved demodulation performance, e.g., by up to 3 dB.

FIG. 15 shows illustrative outputs of the correlator. Particularly, thetop chart 1502 is an overlay of the higher-amplitude analog correlatoroutput (data 1-0), and the lower-amplitude line represents the binaryversion. The second chart 1504 is of an exemplary correlator magnitudesignal.

FIG. 16 is a system diagram of one windowed integrator 1402 of FIG. 14.Note that any type of windowed integrator or suitable alternative knownin the art may be used, and the circuit shown is presented by way ofexample only to show one possible implementation.

FIG. 17 is a system diagram of one variable delay module 1404 of FIG.14. Note that any type of variable delay module or suitable alternativeknown in the art may be used, and the circuit shown is presented by wayof example only to show one possible implementation.

Referring again to FIG. 4, the sliced data output from the correlatormodule 414 may be filtered by a filter module 416, the output of whichis received by a preamble delimiter detection module 418. FIG. 18 is asystem diagram of the preamble delimiter detection module 418 of FIG. 4.Note that any type of preamble delimiter detection module or suitablealternative known in the art may be used, and the circuit shown ispresented by way of example only to show one possible implementation.When processing the incoming signal, the carrier wave is received duringthe carrier preamble period. Then, the preamble delimiter detectionmodule 418 detects a delimiter, e.g., 1-0-1-1-1, which denotes the startof a frame. As shown, the illustrative preamble delimiter detectionmodule 418 includes a section 1802 that counts the time betweentransitions, the output of which is received by a state machine 1804,which is triggered at each transition. The output of the state machine1804 is received by a state sequence recognition module 1806. The statesequence recognition module 1806 generates a preamble delimiterdetection signal, which may be further processed before leaving thepreamble delimiter detection module 418.

As an option, the preamble delimiter detection module 418 may beactivated by an acquisition enable signal (acq_en) output by thederotator 402, as shown in FIG. 4.

FIG. 19 is a system diagram of the state machine 1804 of FIG. 18. Notethat any type of state machine or suitable alternative known in the artmay be used, and the circuit shown is presented by way of example onlyto show one possible implementation.

FIG. 20 is a system diagram of the state sequence recognition module1806 of FIG. 18. Note that any type of state sequence recognition moduleor suitable alternative known in the art may be used, and the circuitshown is presented by way of example only to show one possibleimplementation.

FIG. 21 is a system diagram of the clock recovery module 420 of FIG. 4.Note that any type of clock recovery module or suitable alternativeknown in the art may be used, and the circuit shown is presented by wayof example only to show one possible implementation. As shown in FIGS. 4and 21, the illustrative clock recovery module 420 receives as input thefiltered baseband data from the filter 416, the preamble delimitersignal from the preamble delimiter detection module 418, and thefrequency offset signal from the derotator 402. A frequency jam signalis generated by the jam set section 2102, which converts the estimatedsubcarrier frequency offset signal generated in the correlator 402 (FIG.4, 5) into a scaled version for the clock frequency jam set.

The frequency jam signal is input to a loop filter 2104 as the initialdata clock frequency error.

FIG. 15 shows an illustrative output 1506 of the clock recovery module420. Particularly, in the clock recovery loop, there is a term calledthe clock filter integral. When the preamble delimiter is detected, thedelimiter detect signal is set high, a frequency jam set value derivedfrom the subcarrier recovery section is loaded into the clock filterintegral register, and the clock recovery loop then tracks the clocktiming until the end of the signal, then starts to diverge again.

FIG. 22 is a system diagram of the loop filter 2104 of FIG. 21. Notethat any type of loop filter or suitable alternative known in the artmay be used, and the circuit shown is presented by way of example onlyto show one possible implementation.

Referring again to FIG. 4, an end of modulation detection module 422 maybe used to determine when the modulated signal has ended. FIG. 23 is asystem diagram of the end of modulation detection module 422 of FIG. 4.Note that any type of end of modulation detection module or suitablealternative known in the art may be used, and the circuit shown ispresented by way of example only to show one possible implementation.The end of modulation signal is derived from the correlator magnitudeoutput of the correlator.

The end of modulation signal Output from the end of modulation detectionmodule 422 is used to indicate the end of modulation. This isparticularly useful in situations where the amount of data is unknown,such as where a reader tells a tag to backscatter all contents of itsdata up to the end of a bank, and the reader does not know how many bitsthe tag has. This module 422 provides a way for the system to know whento stop reading and when it can go on to next operation, e.g., startsending data again. In the end of modulation detection circuit 422, thefast averager 2302 may be a 3-point window integrator and the slowaverager 2304 may be a leaky integrator.

FIG. 24 is a system diagram of a 3-point window integrator 2302 of FIG.23. Note that any type of a fast averager or suitable alternative knownin the art may be used, and the circuit shown is presented by way ofexample only to show one possible implementation.

FIG. 25 is a system diagram of the leaky integrator 2304 of FIG. 23.Note that any type of a slow averager or suitable alternative known inthe art may be used, and the circuit shown is presented by way ofexample only to show one possible implementation.

FIG. 26 is a system diagram of the automatic gain control (AGC) module424 of FIG. 4. Note that any type of AGC or suitable alternative knownin the art may be used, and the circuit shown is presented by way ofexample only to show one possible implementation. The AGC forms part ofa loop that attempts to get to the closest coarse gain setting as it canin the preamble period. Once the AGC finishes at the end of preambleperiod, the shift value that the AGC calculates is frozen, which createsa gain setting for the signal.

FIG. 27 is a system diagram of the received signal strength indicator(RSSI) module 426 of FIG. 4. Note that any type of RSSI circuit orsuitable alternative known in the art may be used, and the circuit shownis presented by way of example only to show one possible implementation.The shift signal from the AGC is received by the RSSI module 426. Forevery bit of this shift, a certain amount of gain is applied. Forexample, the range of shift values may be from 1 to 15, and for everybit of shift, there is a 6 dB difference in signal level. To that coarseRSSI value is added a fine RSSI value derived from the correlatormagnitude.

There has thus been described a demodulator circuit according to variousembodiments of the present invention. Again, this has been done by wayof example only. The following aspects may be implemented in thedemodulator circuit described, or a different circuit without strayingfrom the spirit and scope of the present invention.

Any of the modules or portions thereof may be implemented in an FPGA inalternate embodiments.

Moreover, the various signals may be filtered, amplified, etc. andappropriate hardware may be provided for the same.

Subcarrier Frequency Acquisition and Complex Derotation to Baseband

FIG. 28 depicts a process flow for a method 2800 for demodulating aradio frequency signal, according to one general embodiment. Suchmethodology 2800 may be implemented in logic such as hardware, software,combinations thereof, etc. Moreover, the method 2800 may be performed inconjunction with any of the illustrative systems and/or components notedherein, with other types of systems and/or logic as would be known andunderstood by one skilled in the art upon reading the presentdisclosure, and combinations thereof. By way of example and notlimitation, an illustrative hardware and/or software configuration forperforming the method is presented below.

With continued reference to FIG. 28, digital signals derived from aradio frequency signal are received in operation 2802. The radiofrequency signal may be a backscattered signal. An analog to digitalconverter may be used in the circuitry for converting analog signalsfrom the antenna into the digital signals.

In operation 2804, the digital signals are converted to basebandsignals.

In operation 2806, a frequency error signal is generated using thebaseband signals during an acquisition period. In one approach, thefrequency error signal is generated using outputs of a quadraturecorrelator frequency modulation discriminator and a magnitudenormalization module. In some embodiments, the frequency erroraccumulates in a frequency error filter.

In embodiments where the digital signals include sideband signals, logicfor generating the frequency error signal may be initiated after thesideband signals are detected by a subcarrier detection module.

The acquisition period in one approach is the period during a preambleperiod of an incoming radio frequency signal. The frequency of thedigital signals may be frozen at an end of the acquisition period.

In operation 2808, a frequency of the digital signals is shifted towardszero frequency error during the acquisition period using the frequencyerror signal, optionally with the proviso that the digital signals arenot phase locked during the shifting.

In one approach, a frequency offset corresponding to the shiftedfrequency of the digital signals may be used to jam set a data clockrecovery loop. In another approach, the frequency error signal may beused to control a numerically controlled oscillator, and using an outputof the numerically controlled oscillator to determine multipliers for aderotator that outputs the baseband signals.

A system according to one embodiment includes a multiplier coupled tosignal inputs, the multiplier being for outputting baseband signalsderived from subcarrier signals received via the signal inputs; and afrequency error detector module coupled to baseband signal outputs ofthe multiplier, the frequency detector being for generating a frequencyerror signal using the baseband signals from the multiplier, wherein anoutput for the frequency error signal is coupled to an NCO, wherein anoutput of the NCO is used as a reference input for the multiplier,thereby creating a frequency acquisition loop, also referred to as afrequency error loop.

The system may further include at least one antenna and an analog todigital converter for converting analog signals from the antenna intothe digital signals. In one approach, the frequency detector moduleincludes a quadrature correlator frequency modulation discriminator anda magnitude normalization module.

Referring again to the illustrative embodiment described above withrespect to FIG. 4, the reverse link backscatter link frequency (BLF) insome approaches may vary from 25 KHz to 640 KHz and above. In addition,there is a large frequency error allowed by some RFID standards thatranges from 4% to >20% of the BLF. In C1G2-compliant passiveapplications, the reverse link signal is assumed to be very strong, so acommon technique in the industry is to look for zero crossings in the Ior Q quadrature arms of the demodulator. They either count the zerocrossings or use a simple correlator at the nominal frequency withshortened length to accommodate frequency error.

In one embodiment of the present invention, the system includes aquadrature correlator FM discriminator with magnitude normalizationduring the preamble period to acquire close frequency acquisition. Thepreamble period generally corresponds to the period of time before thestart of a frame, which may be denoted by a preamble delimiter. FIGS. 9and 11 illustrate a quadrature correlator FM discriminator 906 accordingto one embodiment, the output of which is proportional to the frequencyerror and can thus be used to generate a frequency error (freq_error)signal which goes into a frequency error loop.

As noted above, if the incoming signal is large, the frequency errorappears large because it is proportional to the signal amplitude. But ifthe incoming signal is normalized by the amplitude of the signal, thefrequency error does not appear large. Thus, referring again to FIG. 9,the magnitude normalization module 904 provides the scaling thatultimately is used along with the output of the quadrature correlator906 to generate a proper frequency error signal. This design acts as aquadrature correlator FM discriminator with magnitude normalization.Moreover, because tags typically do not have a good clock, this approachallows reading of the signal no matter what state the subcarrier is in,and in spite of noise.

The frequency estimate that is obtained during the preamble period isused, e.g., in the correlator 402 (FIG. 5), to downconvert thesubcarrier to a complex baseband (near zero) where the signal matchedfiltering is performed. Exact frequency acquisition is advantageouslynot required because the complex correlator can handle relatively smallresidual frequency error.

Referring to FIG. 5, the frequency error signal is used by the frequencyerror filter module 507 to generate an NCO input signal that adjusts thefrequency of the NCO 511. The NCO 511 in one approach may be a moduleaccumulator that counts up to a certain number and when it hits thatmodule (number), it resets back to 0. In a preferred approach, ratherthan reset to 0, the NCO subtracts out the modules. Particularly, assumethere is a certain target frequency, say 160 Khz (8 microseconds). Ifthere is 1 megasample per second (1 sample per microsecond), the NCOcounts 0 to 7, then back to 0. The result is a basic frequency that isdetermined by counting 8 clocks (1 megasample gets the NCO close to 160KHz). The frequency error is used to adjust the frequency. For example,say the frequency error is 0. Then the loop operates on a nominalfrequency determined by the NCO loop increment. If there is no error,then the loop will have exactly this frequency. However, if there is anerror, the frequency of the loop is adjusted. For most applications,only 8 phases are needed to get sufficient accuracy for downconversion,especially where the illustrative embodiment shown in FIG. 4 is used.Thus, the NCO only needs to determine when it crosses ⅛, 2/8, ⅜, etc.and then it rolls back around to 0. Thus, referring to FIG. 7, the NCOmay determine a phase and divide it into 8 equal increments. Referringagain to FIG. 5, the output of the NCO goes into a sin/cosine lookupmodule 522, which may use a sin/cosine lookup table to derive valuesbased on the NCO output. The 8 phases thus become multipliers for thederotators 508, 510. Each derotator 508, 510 does a full complexdownconversion using the sin/cos values coming out of sin/cosine lookupmodule 522.

As the system gets closer to the correct frequency, the frequency errorwill approach 0.

The foregoing allows the system to demodulate signals that are veryclose to the noise floor by using a full Tbit length correlator atbaseband (near zero) frequency, where a Tbit is the length of data bit(e.g., if data rate is 4 Kbits/s, one Tbit is 250 microseconds long).

This also allows the demodulator to tune to a continuous range of BLF's.Using a fraction of the frequency reduces signal power, so by using thefull symbol time, the maximum possible signal is utilized, resulting inabout the highest signal to noise ratio possible. Moreover, if more thanone symbol is integrated, error from next symbol is introduced, so onesymbol length is preferred. The foregoing approach allows use of fullTbit length with minimal degradation due to residual frequency error.

To demonstrate, consider the following example. Prior to or during aread period, the DSP (which includes the demodulator) is turned on. Theincoming and digitized sideband signals go into the derotator, e.g.,derotator 402 of FIG. 4. Subcarrier detection with wideband filtering isinitiated, e.g., using the subcarrier detector module 408 of FIG. 4.Outputs of the derotator go into a filter 403 and into a downsampler.The output of the filter 403 is received by a subcarrier detectionmodule 408 and a frequency detector module 410. The subcarrier detectionmodule 408 looks for energy in a wider band. Upon detecting thesubcarrier signal, the frequency is modified as appropriate to minimizethe frequency error.

As shown in FIG. 5, in one approach, the subcarrier detect signal outputby the subcarrier detection module 408 is received by the subcarrierdetect enable module 509. When the subcarrier detect signal is greaterthan a threshold, frequency acquisition is initiated using anacquisition enable signal (acq_en) for a deterministic length, e.g., thepreamble period. For example, the frequency detector module 410 of FIG.4 may be initiated. A frequency lock loop (frequency error loop) isturned on, and the filtered frequency error is used to drive thefrequency error towards zero during an acquisition period, which maycorrespond to the preamble period. In other words, the frequency of theincoming signal is shifted to minimize the frequency error. In oneapproach, the frequency error slowly accumulates in the frequency lockloop during the preamble period. Preferably, average values are used tokeep the error correction moving in the right direction, even in thepresence of noise. The frequency of the signals is frozen at the end ofthe acquisition period, which in some approaches corresponds to the endof the preamble period.

FIG. 29 is a series of charts showing signals according to oneembodiment. The top chart 2902 depicts a subcarrier detect signal. Onceit goes above a threshold (120 in the chart), it stops. The middle chartdepicts the frequency error, which is used to drive the frequencyacquisition. The lower chart shows the acquisition enable signal, whichis a binary signal that goes high when the subcarrier detect signalexceeds the threshold, and stays high for a period of time just short ofthe preamble period to allow the frequency acquisition loop to pull in.The second line is the resulting frequency that the loop is trying tosettle to. As shown, the second line settles out just before theacquisition enable signal goes low.

As noted above, a quadrature downconverter may be used to downconvertthe incoming sideband signals (subcarrier signals) to baseband duringrecovery of the tag's modulation. Previously, devices picked the railwith the strongest signal, then looked for zero crossings on that signalto determine frequency cycles. A problem with this approach, however, isthat if there was any Doppler in the incoming signal, a beat frequencydevelops when the two sidebands in the selected rail are combined. Inother words, a precess through the phase on the I and Q develops becausethe carrier used for the downconversion is no longer as sent out, i.e.,the frequency changes due to the movement of the tag. The result is thedecoding is less reliable. Diverging from the conventional approach, inone embodiment, the DSP receives the signals from both incoming railsand processes both signals, i.e., performs a complex downconversion onsignals from both rails.

The corresponding frequency error may also be used to jam set the clocktiming loop, as described below.

Data Clock Recovery Loop Jam Set Using Subcarrier Frequency Estimate

FIG. 30 depicts a process flow for a method 3000 for jam setting aninitial frequency of a data clock recovery loop, according to onegeneral embodiment. Such methodology 3000 may be implemented in logicsuch as hardware, software, combinations thereof, etc. Moreover, themethod 3000 may be performed in conjunction with any of the illustrativesystems and/or components noted herein, with other types of systemsand/or logic as would be known and understood by one skilled in the artupon reading the present disclosure, and combinations thereof. By way ofexample and not limitation, an illustrative hardware and/or softwareconfiguration for performing the method is presented below.

With continued reference to FIG. 30, a frequency error signal isgenerated in a frequency error detector from sideband signals within abackscattered radio frequency signal. The frequency error accumulates ina frequency error filter coupled to an output of the frequency errordetector. See operation 3002. In one approach, the frequency error isdetected at the baseband output of a complex multiplier in a subcarrierderotator. The subcarrier derotator may include the frequency errordetector and/or the frequency error filter. Complex outputs of an RF tobaseband downconverter may be digitized and input to the complexmultiplier. For example, the frequency error signal may be generatedusing outputs of a quadrature correlator frequency modulationdiscriminator and a magnitude normalization module. As an option, thelogic for generating the frequency error signal is initiated after thesideband signals are detected by a subcarrier detection module.

In operation 3004, at about an end of an acquisition period, theaccumulated frequency error in the frequency error filter is frozen. Theacquisition period may be during a preamble period of the backscatteredradio frequency signal.

In operation 3006, the frozen accumulated frequency error is used to jamset an initial frequency of a data clock recovery loop. In one approach,the data clock recovery loop is jam set upon detection of a preambledelimiter in the backscattered radio frequency signal. In oneembodiment, a scaled version of the frozen accumulated frequency erroris used as an initial “jam set” value of an integral term of a dataclock recovery loop filter.

A system according to one illustrative embodiment includes one or moreof a multiplier coupled to signal inputs, the multiplier being foroutputting baseband signals derived from received signals; a frequencyerror detector module coupled to baseband signal outputs of themultiplier, the frequency error detector being for generating afrequency error signal using the baseband signals from the multiplier,wherein an output for the frequency error signal is coupled to themultiplier thereby creating a frequency error loop; a data clockrecovery loop coupled to the frequency error loop, the data clockrecovery loop having logic for receiving a frequency offset signalcorresponding to a subcarrier frequency offset; and a preamble delimiterdetection module that outputs a preamble delimiter detect signal upondetecting a preamble delimiter in the received signals, wherein the dataclock recovery loop is jam set upon output of the preamble delimiterdetect signal. At least one antenna may be present for receiving thesideband signals. The system may further include logic for digitizingcomplex outputs of an RF to baseband downconverter, the digitizedoutputs being input to the multiplier.

The frequency detector module may include a quadrature correlatorfrequency modulation discriminator and a magnitude normalization module.

In one embodiment, the frequency offset is determined in the frequencyerror loop by shifting a frequency of the digital signals towards zerofrequency error during the acquisition-period using the frequency errorsignal.

In one approach, a scaled version of an accumulated frequency error isused as an initial “jam set” value of an integral term of a loop filterin the data clock recovery loop.

Referring again to the illustrative embodiment described above withrespect to FIG. 4, the data clock recovery loop used in one embodimenttracks both phase and frequency errors. In C1G2 and C3 (and other) tags,both subcarrier frequencies and data clock frequencies are derived offof a single tuned circuit, so the frequency errors in the backscatteredsignal are proportional. This proportionality is true at any combinationof BLF and data rate, which allows the demodulator of variousembodiments of the present invention to tune to a continuous range ofdata rates.

In one illustrative embodiment, the input to the clock recovery moduleconies from the correlator module, where the correlator module outputsdata pulses corresponding to the baseband. Again, if the baseband signalis sampled at the appropriate time, its character as a one or a zero canbe determined. The clock recovery module provides the timing. Once thepreamble delimiter is detected, though the clock loop has not beenrunning, the preamble is a waveform shape such that the start timing ofthe first data bit can be derived therefrom and used to derive an errorestimate that can in turn be used to jam set the clock. Once the clockloop is running, it can begin tracking the error in the clock timing.

If the clock recovery module were not jam-set, a few cycles would haveto occur until enough information has been accumulated to provide acorrect frequency offset. However, the knowledge of the approximatefrequency of the particular tag being communicated with is leveraged toestimate that the clock is off in a proportional amount to thesubcarrier offset.

In one embodiment of the present invention, the data clock recovery loopis a full second order loop that tracks both phase and frequency toobtain a more accurate clock recovery. See, e.g., the data clockrecovery loop in the clock recovery module 420 of FIG. 21. The phase maybe set using the delimiter timing, and the frequency can be set becausethe system has already done a frequency acquisition of the subcarrier,e.g., in module 410 of FIG. 4.

In some approaches, the clock recovery module 420 uses a scaled versionof the frequency error estimate from the subcarrier frequencyacquisition section as the initial data clock frequency error. In otherwords, the frequency error derived from the frequency detector may beused to jam set the data clock recovery loop. Because the system hasdetermined the frequency error, the data clock recovery loop can beinitiated with the same knowledge because the correct frequencyinformation may have been determined even before the clock timing hasstarted. By using a good estimate of the correct offset to start with,the system is more quickly able to track each bit in the incomingsignal.

Referring again to FIG. 4, in one embodiment, the preamble delimiterdetect signal output from the preamble delimiter detection module 418 isreceived by a clock recovery module 420. As mentioned above, whenprocessing the incoming signal, after the preamble period, the systemdetects a delimiter, e.g., 1-0-1-1-1, which denotes the start of aframe. Preferably, the clock recovery module 420 does not turn on untilthe delimiter detect module 418 outputs a detection signal. When thedelimiter detect signal is output, the clock is jam set. Then, anintegrator may be used to generate the clock sync signal. The data clockfrequency is determined and divided to get the data rate.

Thus, the frequency error signal can be used to shift both the dataclock as well as the subcarrier frequency.

Variable Length Correlator

FIG. 31 depicts a process flow for a method 3100 for processing a signalderived from a radio frequency signal at some rate in a range ofallowable data rates, according to one general embodiment. Suchmethodology 3100 may be implemented in logic such as hardware, software,combinations thereof, etc. Moreover, the method 3100 may be performed inconjunction with any of the illustrative systems and/or components notedherein, with other types of systems and/or logic as would be known andunderstood by one skilled in the art upon reading the presentdisclosure, and combinations thereof. By way of example and notlimitation, an illustrative hardware and/or software configuration forperforming the method is presented below.

With continued reference to FIG. 31, an incoming signal derived from aradio frequency signal is downconverted to complex near-basebandsignals. See operation 3102.

In operation 3104, the complex near-baseband signals are processed intwo data correlators, one correlator corresponding to data 0 and theother corresponding to a data 1. In a preferred embodiment, eachcorrelator is a Tbit windowed integrator, wherein Tbit is a duration ofa data symbol and is variable over a predetermined range, and where aresidual subcarrier frequency on the near-baseband signal input to thecorrelator is small relative to the data rate, e.g., less than about 10%of the data rate.

In another embodiment, each correlator includes a series of windowedintegrators that are configurable for different lengths. In one approachoutputs of the windowed integrators of each correlator may be added inone path and subtracted in another path and processed bycomplex-to-magnitude modules to generate the data signals, where thedata signals are kept separate until after outputs of thecomplex-to-magnitude modules. In another approach, each of the windowedintegrators of each correlator is a half symbol integrator correlator,where correlator variable delay modules provide a delay into two of thewindowed integrators, wherein no delay is introduced into another two ofthe windowed integrators.

In operation 3106, the effective lengths of the correlators are changedbased on a symbol data rate of the incoming signal.

In one embodiment, the correlator outputs are combined to create acombined correlator magnitude signal, and the combined correlatormagnitude signal is used to determine an end of a signal modulation.

Referring again to the illustrative embodiment described above withrespect to FIG. 4, signal matched filtering in some embodiments may beaccomplished by first downconverting to complex baseband (near zero),then using a Tbit windowed integrator for the data correlator. Thisallows simple tuning for different data rates, i.e., data bit lengths intime, by changing the length of the correlator to the appropriate numberof samples based on the data rate of the baseband signal.

In particularly preferred embodiments, the correlator acts as a squarewindowed integrator in which if the number of samples being integratedchange, the data time also changes, thereby allowing the system to tuneto different data lengths.

Particularly, after the system acquires a certain number of samples persymbol, no more are needed. But as data rate gets lower, the sample ratestays the same, but bit data times get longer. Thus, the system canignore several of the samples, using only those it needs, e.g., ignores4 out of 8 samples, ignores 7 out of 8 samples, etc.

As shown in FIG. 4, output signals from the derotator 402, in additionto going to the frequency detector module 410 and subcarrier detectormodule 408, also goes to the correlator module 414. Referring to FIG.14, the correlator module 414 includes a series of correlator blocks1402 for positive baseband signals and another series for negativebaseband signals. In the embodiment shown, the correlator blocks 1402are windowed integrators As noted above, the windowed integrators mayeach be configurable for different lengths thereby allowing the systemto handle a large variety of data lengths.

Half Bit Correlator Combining for Data Demodulation

FIG. 32 depicts a process flow for a method 3200 for demodulating a datasignal, according to one general embodiment. Such methodology 3200 maybe implemented in logic such as hardware, software, combinationsthereof, etc. Moreover, the method 3200 may be performed in conjunctionwith any of the illustrative systems and/or components noted herein,with other types of systems and/or logic as would be known andunderstood by one skilled in the art upon reading the presentdisclosure, and combinations thereof. By way of example and notlimitation, an illustrative hardware and/or software configuration forperforming the method is presented below.

With continued reference to FIG. 32, baseband signals are processedusing multiple complex correlators each having a half-Tbit length. Seeoperation 3202. In one approach, each of the complex correlatorsincludes a delayed integrator and a nondelayed integrator. In anotherapproach, each of the complex correlators includes a series of windowedintegrators that are configurable for different lengths. The basebandsignals may be derived from a Miller modulated signal or an FM0modulated signal in some approaches.

In operation 3204, outputs of the complex correlators are summed for afirst data state. In one approach, the summed outputs are kept separateuntil after outputs of complex-to-magnitude modules

In operation 3206, outputs of the complex correlators are differencedfor a second data state.

The method may further include generating a correlator magnitude signal,and using the correlator magnitude signal to determine an end of asignal modulation. In another variation, the method may further includegenerating a correlator magnitude signal, and using the correlatormagnitude signal to adjust an automatic gain control setting. In yetanother embodiment, the method may include generating a correlatormagnitude signal, and using the correlator magnitude signal to derive areceived signal strength indicator (RSSI) estimate.

Referring again to the illustrative embodiment described above withrespect to FIG. 4, two half Tbit correlators may be used to provide bothdata 0 and data 1 correlators with half the logic and using sum anddifference combining at the output of the half Tbit correlators.

The underlying encoding for Miller modulated signals is a data zerowhich does not have a mid-symbol transition and a data one which has amid-symbol transition. FIG. 33 illustrates exemplary data 3302 receivedfrom the tag, and the Miller basis modulation 3304 for the data 3302.The Miller basis modulation from the tag is based on the subcarrier withmodulation on it, where the modulation is generated by output from aMiller generator combined with a BLF frequency generator output. Alsoshown in FIG. 33 is an exemplary carrier wave signal 3306 and anexemplary reset signal 3308. The preamble period of the carrier wavegenerally runs between the time the reset signal goes low and the startof the Miller encoded signal, which may include a preamble delimiter.

To perform the highest performance correlation for both datapossibilities with the least computation, two complex correlators withhalf-Tbit lengths may be used. The outputs of each are combined with asum for a data state corresponding to zero and a difference for a datastate corresponding to one (or vice versa, e.g., for FM0 encoding). Thisallows the system to do both correlations with a single Tbit lengthcorrelator and a complex add and subtract. In other words, thecorrelation for both a data 0 and 1 can be performed using one set ofcorrelators, as opposed to using different sets of correlators for eachstate.

For example, assume the correlator 414 of FIG. 14 is being used. In thewindowed integrator 1402, a sample is received, but also delay by anumber of samples (e.g., 6, 8, 10, 12, 16, etc.). See FIG. 16. Thesignal is integrated, e.g., added. The signal stays in the windowedintegrator for e.g. 6 (or 8 or 10 or 12 or 16) samples. Then after the6th time it is subtracted out or flushed from the windowed integrator.For example, assume a 1 followed by five 0's comes in. The value will be1 until the 7th cycle, where 1 becomes a −1. In another example, assumeall 1's coming in. The value will be 6. In the 7th cycle, assume another1 comes in, and a −1 is applied, so the value stays at 6. The result isthe same as if a 6 bit register were used, but with not near as muchcomplexity as those approaches using a shift register, though these maybe used in some approaches.

Automatic Gain Control and/or Baseband Preintegrator to ReduceComputation Requirements at Lower Data Rates

FIG. 34 depicts a process flow for a method 3400 for processing basebandsignals, according to one general embodiment. Such methodology 3400 maybe implemented in logic such as hardware, software, combinationsthereof, etc. Moreover, the method 3400 may be performed in conjunctionwith any of the illustrative systems and/or components noted herein,with other types of systems and/or logic as would be known andunderstood by one skilled in the art upon reading the presentdisclosure, and combinations thereof. By way of example and notlimitation, an illustrative hardware and/or software configuration forperforming the method is presented below.

With continued reference to FIG. 34, I and Q baseband signals arereceived in operation 3402.

In operation 3404, an amount of samples of the baseband signals to beprocessed in a correlator is selectively reduced, where the reductionrate is based on a data rate of the baseband signals (e.g., 2 to 1, 4 to1, 8 to 1, etc.). For example, the amount of samples may not be reduceduntil a data rate of the baseband signals falls below a threshold. Theamount of output samples may be reduced using any type of scheme. In oneapproach, the amount of output samples is reduced by preintegratinginput samples by adding a selectable number of the input samplestogether.

In an optional operation, an automatic gain control may be performedprior to reducing the amount of samples to reduce a number of bitsprocessed in the reducing step.

A preintegrator module according to one embodiment includes an automaticgain control section for performing automatic gain control on I and Qbaseband signals. A first preintegrator is coupled to an output of theautomatic gain control section, the first preintegrator being forselectively reducing an amount of samples in the I baseband signal basedon a data rate of the I baseband signal. A second preintegrator is alsocoupled to an output of the automatic gain control section, the secondpreintegrator being for selectively reducing an amount of samples in theQ baseband signal based on a data rate of the Q baseband signal.

The demodulator in some embodiments uses a variable preintegrator toreduce an amount of samples when the data rate drops below a thresholdrate. At the highest data rates, there is minimal oversampling. As thedata rates drop, the amount of oversampling increases, which improvesclock recovery and demodulation. However, above a certain threshold ofoversampling which corresponds to lower data rates, there is noimprovement to performance; rather the correlator delays get longerwhich would require more area and more computation. Thus, an amount ofsamples in the baseband signals may be selectively reduced based on thedata rate of the baseband signals. Using a preintegrator prior to thecorrelator to limit the amount of oversampling and delay lengthsconstrains processing resources required while still allowing thedemodulation of a wide range of data rates.

Referring again to the illustrative embodiment described above withrespect to FIG. 4, as noted previously, after a certain number ofsamples are received, it generally does not help to process moresamples, and moreover could require a larger number of registers or thelike in the design. Thus, when a certain number of samples is reached,the system may preintegrate by 2, 4, 8, 16, etc. samples. Thepreintegrator module 412 shown in FIG. 12 adds a selectable number ofsamples together. This acts a filtering mechanism. The preintegratormodule 412 also does a scaling to get back to a regular value. Thepreintegrator allows the system to handle a wide variety of data rates,e.g., 1.25 Kbits/s to 160 Kbits/s.

In one approach, the amount of samples is not reduced until a data rateof the baseband signals falls below a threshold.

Single or Dual Complex Subcarrier Downconversion and Data Detection

FIG. 35 depicts a process flow for a method 3500 for subcarrierdownconversion and data detection, according to one general embodiment.Such methodology 3500 may be implemented in logic such as hardware,software, combinations thereof, etc. Moreover, the method 3500 may beperformed in conjunction with any of the illustrative systems and/orcomponents noted herein, with other types of systems and/or logic aswould be known and understood by one skilled in the art upon reading thepresent disclosure, and combinations thereof. By way of example and notlimitation, an illustrative hardware and/or software configuration forperforming the method is presented below.

With continued reference to FIG. 35, a determination is made as towhether to use an upper sideband frequency section of a complexdownconverter output, a lower sideband frequency section of the complexdownconverter output, or both sideband frequency sections of the complexdownconverter output. See operation 3502. In one approach, thedetermination to use the output of the upper sideband section or thelower sideband section is made to avoid interference. In anotherapproach, a determination to use the outputs of both the upper sidebandsection and the lower sideband section is made to improve thesensitivity, e.g., to allow recovery of lower signal strengths thancould be recovered using only one of the outputs.

In operation 3504, the output corresponding to the selected sidebandfrequency section or sections is processed based on the determination,where the processing includes correlation and data detection.

In one embodiment, the determination may be to use outputs of bothsideband sections, where each sideband section includes a correlator,and where the outputs of the correlators are combined after magnitudedetection and before data slicing. In one approach, each correlatorincludes a series of windowed integrators that are configurable fordifferent lengths. Further, each of the windowed integrators may be ahalf symbol integrator correlator. As an option, correlator variabledelay modules may provide a delay into one, two or more of the windowedintegrators, while no delay is introduced into another one, two, or moreof the windowed integrators.

Referring again to the illustrative embodiment described above withrespect to FIG. 4, the complex subcarrier downconverter in someembodiments can be selected to use either the upper or lower subcarriersideband to avoid interference, or it can use and coherently combineboth sidebands while avoiding any zero beat distortion caused by Dopplerfrequency shifts from moving tags. Combining both sidebands in acoherent manner after the correlator improves demodulation performanceby, e.g., up to 3 dB, while avoiding the zero beat problem.

In another embodiment, two derotators may be provided, e.g., one thatderotates the positive subcarrier signal down and another that derotatesthe negative subcarrier signal up. The correlator module 414 of FIGS. 4and 14 has two complete paths, one for each of the positive and negativesubcarrier baseband signals from the derotator modules, which may havebeen scaled in the preintegrator module prior to reaching the correlatormodule. The outputs of the paths are combined after the complex tomagnitude conversion. This dual path arrangement adds about 2-3 dB ofsensitivity. The circuitry after the outputs of the correlator modulemay be as otherwise described herein, and need not be duplicated.

Variable Rate Preamble Delimiter Detection Using Ratiometric Methods

FIG. 36 depicts a process flow for a method 3600 for detecting a patternin a signal, according to one general embodiment. Such methodology 3600may be implemented in logic such as hardware, software, combinationsthereof, etc. Moreover, the method 3600 may be performed in conjunctionwith any of the illustrative systems and/or components noted herein,with other types of systems and/or logic as would be known andunderstood by one skilled in the art upon reading the presentdisclosure, and combinations thereof. By way of example and notlimitation, an illustrative hardware and/or software configuration forperforming the method is presented below.

With continued reference to FIG. 36, a time between symbol transitionsin a signal derived from a radio frequency signal is determined. Seeoperation 3602. As an option, a sign of each ratio may be determinedbased on a direction of the transition.

In operation 3604, ratios of relational times between consecutive symboltransitions are determined.

In operation 3606, a sequence of the ratios is compared to a targetpattern for determining whether the sequence corresponds to the targetpattern. In one approach, the target pattern corresponds to a preambledelimiter in a signal derived from a radio frequency signal.

The comparing the sequence of the ratios to the target pattern in oneembodiment includes assigning a value to each ratio based on the ratiofalling within a range corresponding to the value, and comparing aresulting sequence of values to the target pattern. In one approach,each ratio corresponds to more than one value. The target pattern mayinclude a plurality of allowable values for each position in thesequence.

In one embodiment, the method 3600 is performed without input from aclock recovery section. In another embodiment, a detect signal is outputupon determining that the sequence corresponds to the target pattern.The detect signal may be used, for example, to indicate a time to jamset a clock section.

The method 3600 may be implemented in any type of system, such as aradio frequency identification reader and/or an RFID tag.

The preamble delimiter pattern is preferably detected prior to dataclock recovery to establish the framing of the signal data bits and ofthe entire packet of data. Where preamble delimiter detection is doneprior to data clock recovery, the delimiter detector is preferablycapable of detecting the pattern with large frequency errors without theaid of the clock recovery loop. This is accomplished in one embodimentby using a ratiometric method which determines the time between eachunderlying symbol transition (e.g., zero crossing), determines theratios of relational (e.g., consecutive) times between symboltransitions, and matches the sequence of ratios to a target pattern.

At the beginning of a transaction, the system does not know much aboutthe incoming signal: the clock is not yet running; the system does notknow where the start of frame is, where the edge of each symbol is; etc.Moreover, the tag can have a high (e.g., 4%) frequency error. Inaddition, a preferred embodiment of the system can handle two orders ofmagnitude of data rates. All of this would be expected to make it verydifficult to detect the particular pattern that occurs at or in thepreamble delimiter.

Referring again to the illustrative embodiment described above withrespect to FIG. 4, to overcome these obstacles, the system according tosome embodiments looks at ratios. In one approach, the system takestiming measurements, generates ratios of adjacent timings, and looks fora particular pattern of ratios. To illustrate assume that the output ofthe correlator 414 of FIG. 4 goes into a slicer and filter 416, and thefiltered data looks like the pattern 3701 shown in FIG. 37. Timingmeasurements are made between transitions, and ratios of the timingmeasurements are created. As shown in this example, the timemeasurements between transitions for the sample 3701 shown in FIG. 37are 1, 1.5, 1, 2, 1, and 1.5. The ratios of adjacent measurements in are+1.5/1, −1/1.5, +2/1, −1/2, and +1.5/1, which correlate to +1.5, −2/3,+2, −1/2, and +1.5, and where the optional +/− is dependent upon thetransition direction. If this pattern of ratios matches the preambledelimiter pattern, the preamble delimiter detection module outputs asignal indicating that the preamble delimiter has been detected.

Because of noise, the system may not get an exact pattern of ratios,even though it should have. Therefore, it may be preferable to createranges into which the ratios may be expected to fall. Then a value maybe assigned to each ratio based on the ratio falling within a rangecorresponding to the value. For example, a table 3802 of values may beused, as in FIG. 38. If a ratio falls into one of the ranges, the value(e.g., A, B, C, etc.) associated with that range is assigned to theratio. Then, the sequence, e.g., of ABCs is compared to the targetpreamble delimiter pattern to determine if they correspond to eachother, e.g., the sequence of ABCs matches the target pattern. In theexample of FIG. 37, the pattern from the incoming signal is +B−D+A−E+B.Assuming this sequence is the delimiter pattern, the preamble delimiterdetector module outputs a signal denoting that the preamble delimiterpattern has been found.

The system may also use a fuzzy logic approach in the pattern matchingto improve the probability of detection, especially in low signal tonoise ratio (SNR) conditions. In such approach, a given ratio maycorrespond to more than one value, and so any of the potential valuesthat match the appropriate position in the delimiter pattern would beused to determine a match.

In one illustrative embodiment using fuzzy logic, assume the relationaltable 3902 is as shown in FIG. 39. For example the ratio 1.5 from theexample of FIG. 37 can be B, C or D. Assuming the first value of thedelimiter pattern is D, then the first value would be considered amatch.

The fuzzy boundaries allow for noise in the signal. Moreover, where the+/− signs are determined and used, the signs may be definitive.

This fuzzy logic ratiometric approach may enable delimiter detection atall rates without needing any configuration parameters to handle thevarious rates.

FIG. 18 shows a preamble delimiter detection circuit that can be used inone embodiment.

Note that the variable rate preamble delimiter detection usingratiometric methods may be implemented in an RFID tag, as well as in areader. Similar logic and/or circuitry may be used in the tag.

Moreover, the foregoing logic and/or methodology may be used to detectany kind of pattern (e.g., delimiter), even in the presence of largefrequency errors and without input from the clock recovery section,e.g., clock recovery loop.

End of Modulation Detection

The end of signal modulation is ideally detected in various situations,such as where a read command does not specify the amount of words toread but instead commands the tag to backscatter all the memory contentsfrom a pointer to the end of the memory bank. The reader does not knowwhen the message is complete and so continues to accept bits until toldthat the message is done. By determining when modulation ends, thesystem knows it can go on to other operations (e.g., communicating withanother tag). No a priori knowledge of the number of bits a tag willsend is needed.

The end of signal detector according to one embodiment uses a ratio offast vs. slow averages of the signal magnitude. This is independent ofdata rate and BLF and works down to very low SNRs. In operation, thefast average goes up and down faster than the slow average. Based on themoving relationship of the fast and slow averages, the system candiscern the end of modulation.

FIG. 40 depicts a process flow for a method 4000 for detecting an end ofmodulation in a backscattered radio frequency signal, according to onegeneral embodiment. Such methodology 4000 may be implemented in logicsuch as hardware, software, combinations thereof, etc. Moreover, themethod 4000 may be performed in conjunction with any of the illustrativesystems and/or components noted herein, with other types of systemsand/or logic as would be known and understood by one skilled in the artupon reading the present disclosure, and combinations thereof. By way ofexample and not limitation, an illustrative hardware and/or softwareconfiguration for performing the method is presented below.

With continued reference to FIG. 40, a fast average of a magnitude orpower of an incoming signal is generated in operation 4002. In oneapproach, the incoming signal is a correlator magnitude signal output bya correlator module. Magnitude may be some value derived from the signalpower, such as a square root of the power, etc. A window integrator isused to generate the fast average in one approach.

In operation 4004, a slow average of the magnitude or power of theincoming signal is generated. A leaky integrator is used to generate theslow average in one approach.

In operation 4006, an end of modulation is determined based on arelationship between the fast and slow averages. In one approach, thefast and slow averages are moving averages, and the end of modulation isdetermined upon the fast moving average crossing below the slow movingaverage.

In operation 4008, an end of modulation signal is output upondetermining the end of modulation.

FIG. 23 shows an end of modulation detection circuit 422 that may beused in one embodiment. As shown, a correlator magnitude signal isreceived from the correlator module 414 (FIG. 4). The correlatormagnitude signal generally represents the magnitude of the incomingdata, and tends to settle to a fairly consistent level across thesignal.

Two different filters are applied to the correlator magnitude signal:one responding faster than the other. Preferably, the signals are fastand slow averages of the correlator magnitude signal. Because the fastmoving average goes up faster, upon the start of modulation, it willmove up faster and remain higher than the slow average during themodulation period. When the modulation stops, the fast moving averagewill come down more quickly than the slow moving average. When the fastmoving average crosses the slow moving average, the end of modulationsignal goes high (or low), indicating the end of modulation. In the endof modulation detection circuit 422, the fast averager 2302 may be a3-point window integrator and the slow averager 2304 may be a leakyintegrator.

Received Signal Strength Indicator (RSSI) Estimate on Variable LengthCorrelator Output

Information on the strength of the signal returning from the tag hasmany uses including indication of distance and debugging. The RSSI usesthe output of the variable length correlator to obtain the bestsensitivity for measuring the signal level at all data rates.

FIG. 41 depicts a process flow for a method 4100 for estimating astrength of a radio frequency signal, according to one generalembodiment. Such methodology 4100 may be implemented in logic such ashardware, software, combinations thereof, etc. Moreover, the method 4100may be performed in conjunction with any of the illustrative systemsand/or components noted herein, with other types of systems and/or logicas would be known and understood by one skilled in the art upon readingthe present disclosure, and combinations thereof. By way of example andnot limitation, an illustrative hardware and/or software configurationfor performing the method is presented below.

With continued reference to FIG. 41, a correlator magnitude signal isreceived from a correlator module in operation 4102. In one approach,the correlator module has a variable effective length which ischangeable based on a data rate of the signal. The variable effectivelength may affect an offset used to generate the output indicative of astrength of the signal.

The correlator may receive output from a preintegrator, where thepreintegrator has a variable effective length which is changeable basedon a data rate of the signal. The variable effective length may affectan offset used to generate the output indicative of a strength of thesignal.

In operation 4104, a gain scaling is performed on the correlatormagnitude signal.

In operation 4106, the scaled correlator magnitude signal or valueassociated therewith is averaged to generate an output indicative of astrength of the signal.

In optional operation 4108, a coarse strength level may be set based ona shift signal from an automatic gain control module to sum with theaveraged scaled correlator magnitude signal.

In one mode of use, the output indicative of a strength of the signalmay be used to estimate a distance between an antenna and a tag, e.g.,by comparing the output to a table or scale correlating strength todistance, performing a calculation using the output, etc.

In another mode of use, the output indicative of a strength of thesignal may be used to estimate a density of an object between an antennaand a tag, e.g., by comparing the output to a table or scale correlatingthe strength to density, performing a calculation using the output, etc.

Referring again to the illustrative embodiment described above withrespect to FIG. 4, as noted above, an AGC attempts to determine a coarsegain setting in one embodiment. On top of that, the system may perform afine RSSI estimate which is based on a loop. The RSSI module 426 (FIGS.4 and 27) receives the correlator magnitude signal, and does a gainscaling on it in a gain scaling section 2702, depending on what thelevel of the signal is. The goal is to minimize it to a fairly narrowrange. The signal then goes into a lookup table 2704, which may be a logfunction (e.g. 10, 20, etc. log function). This limits the number oflookup elements needed. The signal then goes into a loop 2706 whichintegrates and averages the RSSI, producing a very fine estimate of theRSSI because of the long filtering (averaging) performed on it.

FIG. 15 shows an illustrative output of the RSSI module 424 in thebottom chart 1508. As shown, the RSSI signal comes to a close estimatewith the coarse set, then tracks to get a good estimate of RSSI.

The RSSI output can be used for a variety of purposes. For example, itis indicative of how strong the signal coming in is, which can in turnbe used to determine how difficult it will be to get the signal. If tagis far away but has line of sight to the system, the RSSI output can beused to determine how far away the tag is. If the tag is behindsomething so the RF signal has to travel through the object, the RSSIoutput can be used to determine how dense the object is. (The amount ofabsorption of the signal through the object in and back out affects thesignal strength.)

By basing the RSSI output off of the correlator magnitude signal, thesystem obtains the full benefit of the filtering all the way down to thedata rate. Thus, if the system is using the narrowest data rate, e.g.,1.25 kHz, filtering is obtained down to that point. Thus, the RSSImodule 426 is able to produce an accurate receive signal strengthindication all the way down to sensitivity at the lowest data rates.

In an alternate method, the system performs RSSI on the input signal.However, because there is no filtering on it, it is difficult todetermine signal strength below a certain level. Moreover, the firstmethod is about 20 dB more sensitive.

Dynamic Signal Detection Threshold

The receive noise level at the A/D inputs vary significantly with anumber of factors: antenna return loss, hop channel center frequency(some frequencies have higher noise levels than others), carriercancellation effectiveness, interference, etc. Additional factors mayinclude the transmit power, the amount of reflection from the antennagenerally corresponding to its reflection coefficient, the amount ofreflection from objects in front of the antenna that reflect signalback. Such noise appears in the subcarrier band from which the signal isexpected. This makes effective signal detection in the subcarrierdetector module 408 (FIG. 4) very difficult, as it is looking for energyin a certain bandwidth about the subcarrier. Energy outside thisbandwidth can be filtered to some extent, making it less likely toobtain a false detection. However, within this bandwidth, which includesthe backscatter frequency, the subcarrier detector module looks for someenergy. The detection threshold has to be above the noise floor.However, the problem is that the noise floor varies with a number offactors, including those above.

The best signal detection threshold is one just above the noise floor.Because the noise level changes from hop to hop and with interference,the threshold in one embodiment is adjusted dynamically to get the bestpossible sensitivity at any given time. To enable this, the systemobtains a measure of the noise floor and sets the signal detectionthreshold just above the noise floor. The setting of the signaldetection threshold may be performed in real time (dynamically), oncebefore the transaction (e.g., during the preamble period, after eachfrequency hop, etc.), periodically or continuously during acommunication, etc.

FIG. 42 depicts a process flow for a method 4200 for setting a signaldetection threshold, according to one general embodiment. Suchmethodology 4200 may be implemented in logic such as hardware, software,combinations thereof, etc. Moreover, the method 4200 may be performed inconjunction with any of the illustrative systems and/or components notedherein, with other types of systems and/or logic as would be known andunderstood by one skilled in the art upon reading the presentdisclosure, and combinations thereof.

With continued reference to FIG. 42, a measure of a noise floor in asignal derived from a radio frequency signal received by an antenna isdetermined using a same circuit used to detect a subcarrier signalduring transmitting and prior to sending a command to a transponder torespond. See operation 4202. This ensures that the tag will not betalking, so that the noise can be determined.

In operation 4204, a signal detection threshold is set above the noisefloor.

Operations 4202 and 4204 may be performed continuously, or at anydesired time, such as after each frequency hop, periodically, etc.

In optional operation 4206, a subcarrier signal in the signal isdetected based on the signal detection threshold, and a subcarrier powerdetect signal is output. The subcarrier signal may be derived, e.g.,from a sideband portion of a backscattered signal. In one approach, asubcarrier detect module detects the subcarrier signal and outputs thesubcarrier power detect signal. An illustrative subcarrier detect moduleincludes: a scaler module for scaling baseband signals; a complex tomagnitude module receiving output from the scaler; a leaky integratorreceiving output from the complex to magnitude module; and a subcarrierpower detect signal output coupled to the leaky integrator.

To enhance the method 4200, carrier cancellation of any type, e.g.,known in the art, may be performed prior to determining the measure ofthe noise floor. In such case, the residual noise is measured.

While much of the foregoing has been described in terms of use with RFIDsystems, it is again stressed that the various embodiments may be usedin conjunction with other types of RF devices, such as receive-only RFdevices, 1- and 2-way radios, boards and/or circuits for RF devices,etc.

The present description is presented to enable any person skilled in theart to make and use the invention and is provided in the context ofparticular applications of the invention and their requirements. Variousmodifications to the disclosed embodiments will be readily apparent tothose skilled in the art and the general principles defined herein maybe applied to other embodiments and applications without departing fromthe spirit and scope of the present invention. Thus, the presentinvention is not intended to be limited to the embodiments shown, but isto be accorded the widest scope consistent with the principles andfeatures disclosed herein.

In particular, various embodiments discussed herein are implementedusing the Internet as a means of communicating among a plurality ofdiscrete systems. One skilled in the art will recognize that the presentinvention is not limited to the use of the Internet as a communicationmedium and that alternative methods of the invention may accommodate theuse of a private intranet, a LAN, a WAN, a PSTN or other means ofcommunication. In addition, various combinations of wired, wireless(e.g., radio frequency) and optical communication links may be utilized.

The program environment in which a present embodiment of the inventionmay be executed illustratively incorporates one or more general-purposecomputers or special-purpose devices such facsimile machines andhand-held computers. Details of such devices (e.g., processor, memory,data storage, input and output devices) are well known and are omittedfor the sake of clarity.

It should also be understood that the techniques presented herein mightbe implemented using a variety of technologies. For example, the methodsdescribed herein may be implemented in software running on a computersystem, and/or implemented in hardware utilizing either a combination ofmicroprocessors or other specially designed application specificintegrated circuits, programmable logic devices, or various combinationsthereof. In particular, methods described herein may be implemented by aseries of computer-executable instructions residing on a storage mediumsuch as a carrier wave, disk drive, or computer-readable medium.Exemplary forms of carrier waves may be electrical, electromagnetic oroptical signals conveying digital data streams along a local network ora publicly accessible network such as the Internet. In addition,although specific embodiments of the invention may employobject-oriented software programming concepts, the invention is not solimited and is easily adapted to employ other forms of directing theoperation of a computer.

Various embodiments can also be provided in the form of a computerprogram product comprising a computer readable medium having computercode thereon. A computer readable medium can include any medium capableof storing computer code thereon for use by a computer, includingoptical media such as read only and writeable CD and DVD, magneticmemory, semiconductor memory (e.g., FLASH memory and other portablememory cards, etc.), etc. Further, such software can be downloadable orotherwise transferable from one computing device to another via network,wireless link, nonvolatile memory device, etc.

Moreover, any of the devices described herein, including an RFID reader,may be considered a “computer.”

While various embodiments have been described above, it should beunderstood that they have been presented by way of example only, and notlimitation. Thus, the breadth and scope of a preferred embodiment shouldnot be limited by any of the above-described exemplary embodiments, butshould be defined only in accordance with the following claims and theirequivalents.

What is claimed is:
 1. A method for setting a signal detectionthreshold, the method comprising: transmitting an outgoing signal;determining a measure of a noise floor in a signal derived from a radiofrequency signal received by an antenna during the transmitting andprior to sending a command to a transponder to respond, the determiningbeing performed using a same circuit used to detect a subcarrier signal;and setting a signal detection threshold above the noise floor.
 2. Themethod of claim 1, wherein the determining and setting are performedcontinuously.
 3. The method of claim 1, wherein the determining andsetting are performed after each frequency hop.
 4. The method of claim1, further comprising performing carrier cancellation during sending theoutgoing signal and prior to determining the measure of the noise floor.5. The method of claim 1, wherein the determining is performed when nosignal is received from the transponder.
 6. The method of claim 1,wherein noise in the radio frequency signal received by an antenna isprimarily reflections of the outgoing signal.
 7. A method for setting asignal detection threshold, the method comprising: determining a measureof a noise floor in a signal derived from a radio frequency signalreceived by an antenna using a same circuit used to detect a subcarriersignal during transmitting and prior to sending a command to atransponder to respond; setting a signal detection threshold above thenoise floor; and detecting a subcarrier signal in the signal based onthe signal detection threshold, and outputting a subcarrier power detectsignal.
 8. The method of claim 7, wherein the subcarrier signal isderived from a sideband portion of a backscattered signal.
 9. The methodof claim 7, wherein a subcarrier detect module detects the subcarriersignal and outputs the subcarrier power detect signal, wherein thesubcarrier detect module includes: a scaler module for scaling basebandsignals; a complex to magnitude module receiving output from the scaler;a leaky integrator receiving output from the complex to magnitudemodule; and a subcarrier power detect signal output coupled to the leakyintegrator.
 10. A system for setting a signal detection threshold, thesystem comprising: logic configured to cause transmission of an outgoingsignal; logic configured to determine a measure of a noise floor in asignal derived from a radio frequency signal received by an antennaduring transmitting the outgoing signal and prior to sending a commandto a transponder to respond, the determining being performed using asame circuit used to detect a subcarrier signal; and logic configured toset a signal detection threshold above the noise floor.
 11. The systemof claim 10, wherein the logic configured to determine and setting isadapted to operate continuously.
 12. The system of claim 10, wherein thelogic configured to set is adapted to set the signal detection thresholdafter each frequency hop.
 13. The system of claim 10, further comprisinglogic configured to perform carrier cancellation prior to determiningthe measure of the noise floor.
 14. The system of claim 10, furthercomprising at least one antenna for receiving the radio frequencysignal.
 15. The system of claim 10, wherein the logic configured todetermine determines the measure of the noise floor when no signal isreceived from the transponder.
 16. The system of claim 10, wherein noisein the radio frequency signal received by an antenna is primarilyreflections of the outgoing signal.
 17. A system for setting a signaldetection threshold, the system comprising: logic configured todetermine a measure of a noise floor in a signal derived from a radiofrequency signal received by an antenna using a same circuit used todetect a subcarrier signal during transmitting and prior to sending acommand to a transponder to respond; logic configured to set a signaldetection threshold above the noise floor; and logic configured todetect a subcarrier signal in the signal based on the signal detectionthreshold, and logic for outputting a subcarrier power detect signal.18. The system of claim 17, wherein the subcarrier signal is derivedfrom a sideband portion of a backscattered signal.
 19. The system ofclaim 17, wherein a subcarrier detect module detects the subcarriersignal and outputs the subcarrier power detect signal, wherein thesubcarrier detect module includes: a scaler module for scaling basebandsignals; a complex to magnitude module receiving output from the scaler;a leaky integrator receiving output from the complex to magnitudemodule; and a subcarrier power detect signal output coupled to the leakyintegrator.